Architecture for a digital subscriber line analog front end

ABSTRACT

An analog front end for communicating discrete multitone modulated signals on a subscriber line includes a transmit block, a receive block, and a hybrid packaged within a same integrated circuit. Upstream data to be transmitted to the subscriber line is pre-processed to eliminate even images. A power spectral density shaping filter subsequently substantially eliminates undesired energy in the upstream data signal. A high pass filter further rejects upstream data from the downstream data signal. The power spectral density filter and the high pass filter enable the use of a first order hybrid network for extracting the downstream data. The analog front end may include an additional analog channel to enable voiceband (e.g., v.90) communication concurrent with non-voiceband (e.g., xDSL) operation. Sample rate conversion is utilized to avoid the use of multiple independent clocks for otherwise incompatible clocking requirements.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of application Ser. No. 09/847,187filed May 1, 2001.

FIELD OF THE INVENTION

This invention relates to the field of telecommunications. Inparticular, this invention is drawn to combining digital and analogtelecommunications functions.

BACKGROUND OF THE INVENTION

Numerous communication protocol standards have developed to enable usingthe pre-existing plain old telephone system (POTS) infrastructure forcarrying digital data. Although the public switched telephone network(PSTN) is digital in nature, the connection between subscribers and thecentral office which serves as an entry point to the PSTN is analog. Asa result, modems are used for bi-directional communication of digitaldata on the analog channel between the subscriber and the centraloffice. Modems convert the communicated information between the digitaland analog domains in accordance with the particular communicationprotocol.

Some communication protocols are designed to rely on the voicebandregion of the analog channel to convey information. As a result, when asubscriber line is in use by such a voiceband modem, the line isunavailable for simultaneous voice communications.

Digital subscriber lines (xDSL) services can provide significantlyhigher data transmission rates by utilizing communication bandwidthbeyond but excluding the voiceband. As a result, xDSL services maysimultaneously co-exist with voiceband communications.

Modems or other devices designed for communicating digital data on theanalog channel utilize an analog front end for transmitting as well asreceiving information from the subscriber line. The analog front endconditions signals communicated to or from the subscriber line beforeproviding the conditioned signal to the subscriber line for transmissionor to a digital signal processor for interpretation.

A hybrid circuit, for example, is used to address echoes resulting fromusing the same two wires for both transmission and reception on theanalog channel. One disadvantage of typical hybrid designs is that thehigh order introduces distortion and noise into the system.

Preferably, a modem has the ability to support the higher data rates ofxDSL when available. Due to the geographical limitations on xDSL,voiceband modems are still needed to ensure a reliable means ofcommunication. One risk averse solution implements the functionality ofvoiceband and xDSL modems through the use of a chipset mounted on acommon circuit board.

The combination of the circuitry onto a common board introduces newproblems. In particular, the voiceband and xDSL modems rely on clocks ofdifferent frequencies which are not multiples of each other. Theinteraction between two clocks, for example, may result inintermodulation, synchronization problems, or other issues whichinterfere with the digital signal processor's ability to properlyinterpret received information.

SUMMARY OF THE INVENTION

Methods and apparatus incorporated into an analog front end forcommunicating with a subscriber line enable a high degree ofintegration.

Upstream data to be transmitted to a subscriber line is pre-processed toeliminate even images. A power spectral density shaping filtersubsequently substantially eliminates undesired energy in the upstreamdata signal. A tunable hybrid network is coupled to the subscriber lineand the transmit block to extract downstream data from the subscriberline. A high pass filter further rejects upstream data from thedownstream data signal. The power spectral density shaping filter andthe high pass filter enable the use of a tuned low order hybrid networkto achieve adequate near end echo cancellation. The analog front end mayinclude an additional analog channel to enable voiceband (e.g., v.90)communication concurrent with non-voiceband (e.g., xDSL) operation.Sample rate conversion is utilized to avoid the use of multipleindependent clocks for otherwise incompatible clocking requirements.

One embodiment of an analog front end apparatus includes a transmitblock coupled to transmit discrete multitone modulated upstream data toa subscriber line. A hybrid network is coupled to the subscriber lineand the transmit block. A receive block is coupled to the hybrid forreceiving discrete multitone modulated downstream data from thesubscriber line. The transmit block, hybrid network, and receive blockreside within a same integrated circuit package.

Other features and advantages of the present invention will be apparentfrom the accompanying drawings and from the detailed description thatfollows below.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention is illustrated by way of example and notlimitation in the figures of the accompanying drawings, in which likereferences indicate similar elements and in which:

FIG. 1 illustrates the communication spectrum allocated for a subscriberline.

FIG. 2 illustrates an analog front end for communication with asubscriber line.

FIG. 3 illustrates the analog front end transmit block.

FIG. 4 illustrates the analog front end receive block.

FIG. 5 illustrates various hybrid network implementations.

FIG. 6 illustrates one embodiment of a tunable hybrid.

FIG. 7 illustrates a voiceband transmit path with sample rateconversion.

FIG. 8 illustrates a voiceband receive path with sample rate conversion.

FIG. 9 illustrates a table of upsampling and downsampling factors forrealizing various clock rates from a universal conversion rate.

FIG. 10 illustrates one embodiment of a finite impulse response filter.

FIG. 11 illustrates one embodiment of a polyphase pointer and serialinterface pulse generator.

FIG. 12 illustrates one embodiment of a polyphase pointer for thereceive path of FIG. 8.

DETAILED DESCRIPTION

The International Telecommunication Union (ITU) has set forth a seriesof recommendations for subscriber line data transmission. Theserecommendations are directed towards communications using the voicebandportion of the communications spectrum (“V.x” recommendations) as wellas communications utilizing frequency spectrum other than the voicebandportion (e.g., “xDSL” recommendations).

The V.x recommendations have evolved over time to support ever higherdata rates. ITU-T Recs. V.22 bis, V.32, V.32 bis, V.34, and V.90, forexample, relate to increasing bit-per-second data rates of 2400, 9600,14400, 33600, and 56000 bit/sec. Compression standards such as V.42 biscan further increase the effective data rates. Generally, voicebandmodems will use a recommended handshaking protocol to negotiate thehighest possible data rate.

ITU documentation for the more recent recommendations related tovoiceband communications includes “Rec. V.90 (09/98)—A digital modem andanalogue modem pair for use on the Public Switched Telephone Network(PSTN) at data signalling rates of up to 56,000 bit/s downstream and upto 33,600 bit/s upstream,” “Rec. V.34 (02/98)—A modem operating at datasignalling rates of up to 33 600 bit/s for use on the general switchedtelephone network and on leased point-to-point 2-wire telephone-typecircuits,” “Rec. V.32 bis (02/91)—A duplex modem operating at datasignalling rates of up to 14 400 bit/s for use on the general switchedtelephone network and on leased point-to-point 2-wire telephone-typecircuits,” and “Rec. V.32 (03/93)—A family of 2-wire, duplex modemsoperating at data signalling rates of up to 9600 bit/s for use on thegeneral switched telephone network and on leased telephone-typecircuits.” Asymmetric digital subscriber line (ADSL) communicationsrepresent one variant of xDSL communications. Exemplary ADSLspecifications are set forth in “Rec. G. 992.1 (06/99)—Asymmetricdigital subscriber line (ADSL) transceivers” (also referred to as fullrate ADSL), and “Rec. G. 992.2 (06/99) Splitterless asymmetric digitalsubscriber line (ADSL) transceivers” (also referred to as G.LITE).

FIG. 1 illustrates the communication spectrum allocation for asubscriber line. Chart 100 compares the portions of the analog channelused by voiceband modems (POTS 110) as well as xDSL modems (ADSL 130).POTS communications typically use the voiceband range of 300-4000 Hz.ADSL is in a range of approximately 25-1100 KHz. A guard band 120separates the POTS and ADSL ranges.

There are multiple line coding variations for xDSL. CarrierlessAmplitude Phase (CAP) and Discrete Multi-Tone modulation both use thefundamental techniques of quadrature amplitude modulation (QAM). CAP isa single carrier protocol where the carrier is suppressed beforetransmission and reconstructed at the receiving end. DMT is amulticarrier protocol. FIG. 1 illustrates DMT line coding.

DMT modulation has been established as a standard line code for ADSLcommunication. The available ADSL bandwidth is divided into 256sub-channels. Each sub-channel 134 is associated with a carrier. Thecarriers (also referred to as tones) are spaced 4.3125 KHz apart. Eachsub-channel is modulated using quadrature amplitude modulation (QAM) andcan carry 0-15 bits/Hz. The actual number of bits is allocated dependingupon line conditions. Thus individual sub-channels may be carryingdifferent numbers of bits/Hz. Some sub-channels 136 might not be used atall. ADSL uses some sub-channels 134 for downstream communication andother sub-channels 132 for upstream communication. The upstream anddownstream sub-channels may be separated by another guard band 140.

During initialization DMT measures the signal-to-noise ratio of eachsub-channel to assign a data rate. Generally, greater data rates (i.e.,more bits/Hz) are assigned to the lower sub-channels because signals areattenuated more at higher frequencies. DMT implementations may alsoincorporate rate adaption to monitor the line conditions and dynamicallychange the data rate for sub-channels.

FIG. 2 illustrates an analog front end for communicating information onan analog channel carried by a subscriber line between a subscriber andthe central office. In one embodiment, the analog front end includescircuitry for handling voiceband as well as non-voiceband (i.e., xDSL)communications. The analog front end conditions signals communicatedbetween the subscriber line 290 and digital signal processors 270, 280.

In one embodiment, the analog front end 218 (including hybrid 260) isprovided on a single substrate within an integrated circuit package.Within the integrated circuit, non-voiceband communication (e.g., xDSL)is handled by transmit block 230, receive block 240, 2-4 wire block 250,and hybrid 260. The hybrid and 2-4 wire conversion functions may becombined into a common hybrid network block as illustrated. In oneembodiment, the analog front end is implemented as complementary metaloxide semiconductor (CMOS) circuitry within the integrated package. Theintegrated circuit may further comprise additional circuitry to supportvoiceband communications within the same integrated circuit package. Forexample, sample rate converter 210 and codec 212 may be incorporatedonto the same substrate as the xDSL circuitry.

The high level of integration of the analog front end is enabledpredominately by four features. First, in order to ease filtering,digital signal processor 270 eliminates even images of the upstreamsignal through the use of a Fast Fourier Transform (FFT). Second, apower spectral density shaping filter is used to further reduce unwantedimages. These first two features ensure compliance with ITU specifiedtransmit power spectrum density requirements. Third, the combination ofthe double rate FFT and power shaping enable the use of a low orderhybrid rather than a higher order distortion inducing hybrid. The hybridis tunable to permit accommodation of varying line conditions anddownstream distortion. Fourth, a high pass filter following the hybridprovides additional rejection of the transmit signal in the receivepath.

Digital signal processor (DSP) 270 provides information in digital formto the transmit block 230 for communication on the analog channel of thesubscriber line. FIG. 3 illustrates transmit block 230 in furtherdetail.

The transmit path is digital in nature from the digital signal processoruntil the digital-to-analog converter 360. Data from the DSP is providedto the programmable gain amplifier 310 of the transmit block.Programmable gain amplifier 310 provides amplification in the range of 0to -3 dB. Attenuation may be used to adjust the power level, if desired.

The xDSL line coding requires orthogonality between carriers.Orthogonality may be achieved through the use of DSP FFT algorithms. Inone embodiment, the xDSL circuitry supports an ADSL protocol with a 138KHz (32 tones) upstream data rate. Although the Nyquist samplingfrequency for a 138 KHz upstream signal is 276 KHz, DSP 270 performs aFast Fourier Transform at twice the necessary sampling rate (i.e., 552KHz or 128 points instead of the 64 points typically used to process the32 tones) to eliminate the even images of the upstream data and to forcethe spectrum content to zero for frequencies between 142 KHz and 276KHz. Thus in one embodiment, DSP 270 is providing upstream data valuesto the transmit block at 552 KHz.

A high pass filter 320 is provided to reduce spectral leakage at lowfrequencies caused by DSP FFT algorithms. In one embodiment, high passfilter 320 is a third order Butterworth filter with a corner frequencyof approximately 12 KHz and rejection greater than 28 dB for frequenciesof approximately 4 KHz or less.

Interpolator 330 interpolates the filtered signal from 552 KHz to 1.104MHz for spectral power shaping. In one embodiment, interpolator 330 is a4^(th) order interpolator. The transfer function for interpolator 330 isas follows:${h(z)}_{330} = \left( {\frac{1}{N} \cdot \frac{1 - z^{- N}}{1 - z^{- 1}}} \right)^{4}$The ratio of the interpolator frequencies determines${N\left( {N = \frac{f_{sout}}{f_{\sin}}} \right)}.$Thus, $N = {\frac{1.104\quad{MHZ}}{552\quad{KHZ}} = 2.}$

The interpolated signal is processed by a power spectral density (PSD)mask or shaper 340 to ensure compliance with protocol specifications. Inone embodiment PSD shaper 340 incorporates a 6^(th) order Chebyshevfilter designed to attenuate the signal by approximately 40 dBm/Hz. Inone embodiment, PSD shaper 340 has a corner frequency of approximately140 KHz .

Interpolator 350 interpolates the power shaped signal from 1.104 MHz to35.328 MHz for the purpose of rejecting upstream images. In oneembodiment, interpolator 350 is a 6^(th) order interpolator. Thetransfer function for interpolator 350 is as follows:${h(z)}_{350} = {\left( {\frac{1}{N} \cdot \frac{1 - z^{- N}}{1 - z^{- 1}}} \right)^{6}.}$In this case, $N = {\frac{35.358\quad{MHZ}}{1.104\quad{KHZ}} = 32.}$

Digital-to-analog converter (DAC) 360 generates an analog signal fromthe interpolated signal. The analog signal is provided to low passfilter 370. Due to the lack of even images, high DAC sampling rate, andthe high rejection of the overall transmit path digital filter, low passfilter 370 may be implemented as a first order filter to rejectundesired upstream images. In one embodiment, low pass filter 370 has acorner frequency of approximately 250 KHz. Driver 380 drives theupstream signal onto the subscriber line 290.

Referring to FIG. 2, receive block 240 interfaces with subscriber line290 through hybrid 260. In one embodiment, hybrid 260 is a tunable,first order filter. The hybrid network is discussed in further detailwith respect to FIGS. 5-6.

FIG. 4 illustrates receive block 240 in greater detail. The 4 wiresignal carrying the transmitted signal as well as any receivedcommunications are passed through hybrid 410 to perform echocancellation and to convert the 4 wires to 2 wires carryingpredominately the received communications. A first programmable gainamplifier (PGA 412) permits gain adjustments from −8 to 24 dB.Attenuation may be necessary in some cases due to line conditions. Thesignal is then provided to high pass filter and PGA 414 to furtherreduce upstream communications in the receive path. High pass filteralso permits gain adjustments from approximately 6-12 dB.

In one embodiment, high pass filter 414 is a 3^(rd) order filter with acorner frequency of approximately 140 KHz to provide approximately 20 dbrejection to the upstream signal. The use of the high pass filter andPGA 414 may be preferable to the use of DACs to cancel near-end signals.Traditional DAC cancellation techniques do not account for driver noiseor distortion.

The filtered signal is provided to low pass filter and PGA 416. Low passfilter and PGA 416 permits gain adjustments from approximately 0-6 dB.In one embodiment, low pass filter 416 is a 2^(nd) order filter with acorner frequency of approximately 2 MHz. The primary purpose of low passfilter and PGA 416 is to provide anti-aliasing for analog-to-digitalconverter (ADC) 420. The signal is passed through another programmablegain amplifer 418 to provide for additional level control before beingprocessed by ADC 420.

In one embodiment, ADC is a 3^(rd) order sigma-delta converter.Decimator 430 serves to reduce the truncation noise contributed by ADC420 at higher frequencies due to the sigma-delta effect. The digitalsignal is decimated from 35.328 MHZ to 8.836 MHZ.

The output of decimator 430 is provided to a programmable low passfilter 440. Low pass filter provides outband rejection to prevent theoutband signal from folding back into the baseband signal. In oneembodiment low pass filter 440 is a 6^(th) order filter. After low passfilter 440, the signal is passed to decimator 450 which decimates thesignal from 8.836 to 2.208 MHz (the baseband frequency). The decimatedsignal is processed by high pass filter 460 before the 16 bit 2.208 MHzdata is provided to a DSP. High pass filter 460 serves to reject anyremaining audio (i.e., voiceband) components in the processed signal. Inone embodiment, high pass filter 460 is a 2^(nd) order filter.

FIG. 5 compares a system level view of a hybrid network (550) withanother hybrid (500). The hybrid is designed in part to cancel thesignal X being transmitted onto the subscriber line so that the largetransmit signal is not re-introduced into the receive path. The signal Xtransmitted by the driver 510 is distorted by the transformer and lineimpedance in a manner modeled as H1(s) 520. The receive path sees asuperposition of the desired input signal Y and X·H1(s). H2(s) 522 isdesigned to introduce the same distortion as H1(s) 520. When combined bydifferential summer 530, the result is Y+X·H1(s)−X·H2(s). IfH1(s)=H2(s), the received result is the desired result, i.e., Y.

One disadvantage with this approach is that the desired receive signal,Y, is usually much smaller than X. Differential summer 530 is taking thedifference of two larger values in order to identify a smaller value.This tends to introduce noise into the system. Another disadvantage isthat H2(s) is a pole/zero filter which does not block DC. Furthermore,the pole and zero are difficult to adjust independently in practicalimplementations. An alternative hybrid network is illustrates as 550 forcomparison.

In this case H1(s) 570 modifies X such that the receive path seesY+H1(s)·X. A graphical representation of a typical H1(s) is illustratedin chart 590. The typical H1(s) has a zero at F1 and a pole at F2. Ahigh pass filter 582 with a corner frequency of F2 is inserted into thereceive path to produce HPF2(s)·(Y+H1(s)·X). Another high pass filter572 with a corner frequency F1 is applied to the transmitted signal X toproduce HPF1(s)·X. Differential summer 580 thus produces,HPF2(s)·Y+HPF2(s)·H1(s)·X−HPF1(s)·X.

The corner frequency of high pass filter 572 produces a pole whichoffsets a zero of H1(s) at the same location. The product of bothfilters appears as another high pass filter with a corner frequency ofF1. High pass filter 570 is selected such that HPF2(s)·H1(s)=HPF1(s).Thus the hybrid is able to reject the transmitted signal, X.

One advantage of this approach is that filters 572 and 582 are the sametype of filter. Moreover, filters 572 and 582 are first order filters.Filters 572 and 582 may be realized, for example, with capacitors andresistors. An added benefit is that high pass filters provide DCisolation and some rejection of the voiceband signal.

FIG. 6 illustrates one embodiment of hybrid 260 in greater detail.Hybrid network 620 includes a hybrid input port, a hybrid output port,and a receive path input port. Driver 612 is coupled to provide theupstream data signal to the subscriber line through resistors R_(D). Thedriver is capacitively coupled to the hybrid input for provide theupstream data signal to the hybrid network. The receive path iscapacitively coupled to the subscriber line for receiving the compositesignal including the upstream and downstream data signals. CapacitorsC₄₁, C₄₂, C₄₄, and C₄₅ provide DC isolation for the hybrid's transmitand receive paths.

A first order model approximates the subscriber line impedance, Z(s), asa series-coupled resistor R_(x) and capacitor C_(x). The transferfunction from the driver 610 output to the receiver input has a dominantzero and a dominant pole as follows:${h_{{drv\_ to}{\_ rxin}}(s)} = {\frac{Z(s)}{R_{D} + {Z(s)}} = \frac{1 + {{sC}_{x}R_{x}}}{1 + {{sC}_{x}\left( {R_{x} + {2R_{D}}} \right)}}}$

The transfer function between the hybrid receive and hybrid output is asfollows:${h_{{rxin\_ to}{\_ HYBout}}(s)} = {K_{rx} \cdot \frac{s}{s + {HYB0}}}$such that the composite transfer function between the hybrid receiveinput and the hybrid output is a high pass function as follows:${{h_{{drv\_ to}{\_ rxin}{\_ toHYBout}}(s)} = {K_{rx}{\frac{1 + {{sC}_{x}R_{x}}}{1 + {{sC}_{x}\left( {R_{x} + {2R_{D}}} \right)}} \cdot \frac{s}{s + {HYB0}}}}},$where K_(rx), is the receive path gain (i.e., the hybrid circuit gain)and HYB0 is the pole of the high pass transfer function (i.e., thehybrid zero). HYB0 is used to minimize the effect of the dominant zero(1+sC_(x)R_(x)) due to line impedance. HYB0 is below 140 KHz so thereceive signal is not significantly affected by the hybrid zero. Thereceive path gain, K_(rx), is controlled by the values of capacitors CRand CF as follows: $K_{rx} = \frac{CR}{CF}$

A hybrid path is used to minimize the effect of the remaining dominantpole (1+sC_(x)(R_(x)+2R_(D))). The transfer function from the driveroutput to the hybrid output through the hybrid path is as follows:${h_{{drv\_ to}{\_ HYBin}{\_ to}{\_ HYBout}}(s)} = {K_{HYB} \cdot \frac{s}{s + {HYBP}}}$HYBP is the hybrid pole. K_(HYB) may be adjusted for different receivepath gains (K_(rx)) such that${{K_{HYB} \cdot \frac{s}{s + {HYBP}}} - {K_{rx} \cdot \frac{1 + {{sC}_{x}R_{x}}}{1 + {{sC}_{x}\left( {R_{x} + {2R_{D}}} \right)}} \cdot \frac{s}{s + {{HYB}\quad 0}}}}->0$The hybrid path gain, K_(HYB), is controlled by the values of capacitorsCG and CF as follows: $K_{HYB} = \frac{CG}{CF}$

The hybrid pole and zero are determined from the components of FIG. 6 asfollows:${{HYB}\quad 0} = \frac{C_{41} + C_{45}}{R_{Z} \cdot C_{41} \cdot C_{45}}$${HYBP} = \frac{C_{42} + C_{44}}{R_{P} \cdot C_{42} \cdot C_{44}}$Thus  ${K_{HYB} = {K_{rx} \cdot \frac{1 + {{sC}_{x}R_{x}}}{1 + {{sC}_{x}\left( {R_{x} + {2R_{D}}} \right)}} \cdot \frac{s + {HYBP}}{s + {{HYB}\quad 0}}}}\quad$Substitution indicates KHYB may be calculated as follows:$K_{HYB} = {K_{rx} \cdot \frac{1 + {{sC}_{x}R_{x}}}{1 + {{sC}_{x}\left( {R_{x} + {2R_{D}}} \right)}} \cdot \frac{s + \frac{C_{42} + C_{44}}{R_{P} \cdot C_{42} \cdot C_{44}}}{s + \frac{C_{41} + C_{45}}{R_{Z} \cdot C_{41} \cdot C_{45}}}}$

Hybrid 260 is thus a tunable low order hybrid. HYBO is adjusted tocancel or ameliorate the zero defined by (1+sC_(x)R_(x)). When properlytuned, hybrid 260 is effectively a first order hybrid. HYBP and K_(HYB)are adjusted to cancel the effects of gain K_(rx) and pole1+sC_(x)(R_(x)+2R_(D)). The optimal K_(HYB) is programmed accordingly.Circuitry 620 is a capacitor coupled amplifier which has almost no noisefrom the passive components. The feedback resistors (RF) for 622 serveas DC stabilizing resistors. When the values for RF are large, theresistors will contribute little noise. The first order hybrid may beimplemented on the same substrate as the remainder of the analog frontend. The high pass filter 414 following the hybrid further reduces theremaining driver signal before the receive signal is digitized by ADC420.

A crystal oscillator is used to generate all the necessary clock signalsfor the digital portion of the analog front end. Although there is somechoice in selection of the crystal frequency, the selected crystal mustsupport the DMT 4.3125 KHz channel separation. Accordingly, the selectedcrystal should have a resonant frequency that is a multiple of 4.3125KHz. In one embodiment, the selected crystal is a 35.328 MHz crystal.

Referring to FIG. 2, analog front end 218 can be configured to support avoiceband channel for voiceband modem communications concurrent with thenon-voiceband (i.e., xDSL) communications. The voiceband channelcircuitry may be integrated on the same integrated circuit die as thexDSL circuitry, if a single clock is used for synchronization. Voicebandcommunications, however, have a bandwidth of approximately 4 KHz andnominal baseband sampling rates of 8 KHz. These values are notimmediately compatible with the requirements of the xDSL circuitryhaving 4.3125 KHz bandwidth subchannels. The use of multiple independentclocks may otherwise interfere with the operation of at least one of thexDSL or voiceband circuitry due to intermodulation and other undesirableeffects resulting from multiple clocks.

Referring to FIG. 2, sample rate conversion is used to derive the otherclock frequencies so that multiple independent clocks can be avoided.Synchronous sample rate conversion is used to achieve the various clockrates through combinations of upsampling by an integer factor anddownsampling by an integer factor. Sample rate conversion block 210includes a transmit path and a receive path which are illustrated ingreater detail in FIGS. 7-8. Actual sample rate conversion is handled byblocks 741, 750, and 752 of FIG. 7 and blocks 844, 850, and 860 of FIG.8. The remaining blocks condition the signal before or after sample rateconversion.

In one embodiment, the voiceband portion of the analog front endsupports a number of clock frequencies including those in the setS={7200, 8000, 8229, 8400, 9000, 9600, and 10,286 Hz). A variety ofclock frequencies allows flexibility in negotiating a common baud ratewith another modem. A universal conversion speed (i.e., intermediatefrequency) is selected which can be derived from any of the requiredoriginal rates by integer upsampling and downsampling. The appropriateupsampling and downsampling integers may be programmatically selecteddepending upon what baud rate the user is attempting to communicatewith. Subsequent digital processing is handled at pre-determinedintermediate frequencies independently of the original frequency.

The sampling frequency f_(s) is selected such that${f_{s} = \frac{f_{o}}{n}},$where n is an integer and f₀ is divisible by n such that f₀mod(n)=0. Inone embodiment, n=32 so that$f_{s} = {\frac{35.328\quad{MHz}}{32} = {1.104\quad{{MHz}.}}}$The conversion speed, f_(audio), is selected judiciously such that forevery f_(i) in S there exists integers M_(i) and N_(i) such that$f_{i} = {\frac{M_{i}}{N_{i}}f_{audio}}$

In one embodiment, the selected universal conversion speed is 9600.Table 900 of FIG. 9 illustrates nominal M and N values when convertinganalog signal rates to the same universal conversion speed of 9600 forfurther digital processing. Common multiples of M and N help to reducethe number of distinct Ms or Ns while maintaining the same ratio. Thusinstead of the fully reduced values of column 2, the use of reducibleratios requires only 3 distinct M: 6, 14, and 16. Although a leastcommon multiple could be used to further reduce M, higher values may notbe desirable for N or for digital filters such as finite impulseresponse filter 750. Thus column 3 illustrates the ratios optimized tominimize the number of distinct Ms for small M. The ratios provideprecise conversions except for 10,286 and 8229 which have errors lessthan 0.5 Hz as required by ITU standards.

Referring to the transmit path 700 of FIG. 7, the optimum M and N areselected from column 3 in accordance with the selected analog speed fromTable 900. Thus, for example, if the selected analog speed is 7200, thenM=16 and N=12. Block 752 upsamples the audio signal by M. The upsampledsignal is provided to finite impulse response (FIR) filter 750. Thereduction in the total number of Ms reduces the number of filtersrequired and therefore the memory required for filter coefficients.After digital filtering, the filtered signal is downsampled by N inblock 742. The data is now at the universal conversion speed of 9600.The downsampled signal is passed to compensator 740 to shape thefrequency response before upsampling. Compensator 740 provides frequencycompensation at high frequencies.

In block 732, the compensated signal is upsampled by 5 and then filteredby infinite impulse response filter 730 to eliminate images introducedby the upsampling. The filtered signal is upsampled again by 23 in block722. The upsampled signal is then passed to interpolator 720 toeliminate the images introduce by the second round of upsampling. In oneembodiment, the 23 times upsampling and the interpolation are performedby the same filter. The interpolated signal is then provided to DAC 710.

The frequency corner for the interpolator is selected to ensure adequateattenuation of signal images as well as no aliasing in the decimationprocess. This requires the FIR 750 cut-off frequency to be the smallerof the interpolation and decimation cut-off frequencies.

FIG. 10 illustrates the operation of FIR 750. The FIR comprises aplurality of samples 1000 of the input data. Each sample 1000 is delayedwith respect to its adjacent sample. Each tap has an associatedcoefficient 1010. The products of the coefficients 1010 and theirassociated tap values (from a preceding sample) are computed and thensummed by adder 1020 to produce a result. In the particular instance,every M coefficients 1010 is used to calculate a new output value. Theinput data shifts N places between calculation of output values. Wheninterpolating, a number of the input sample values will have a zerovalue. FIG. 11 illustrates a polyphase pointer and serial interfacepulse generator 1100 for the FIR of the AFE's transmit path.

FIG. 8 illustrates the receive or downstream path 800. ADC 810 samplesat 1.104 MHz. The resulting digital signal is passed to decimator 820.The decimated signal is downsampled by 23 in block 822. In oneembodiment, the functions of downsampling by a factor of 23 anddecimation are performed by the same filter. At this point the data isbeing communicated at 48 KHz. Block 830 scales the data values, ifnecessary, to accommodate computational requirements of filters 840 and850. In one embodiment, block 830 scales the data values by 64. Infiniteimpulse response filter (IIR) 840 is applied to remove the outbandsignals that may be aliased to the baseband range of 9600 Hz to 48 KHz.The resulting data is downsampled by a factor of 5 in block 842. At thispoint the data is a 9600 Hz signal.

M and N are selected again based on the pre-determined analog speed fromTable 900. Block 844 upsamples the signal by N before providing theupsampled signal to FIR filter 850. The signal is then downsampled by Mso that it has the proper rate for the voiceband channel. FIG. 12illustrates one embodiment of a polyphase pointer for the receive pathof FIG. 8.

In the preceding detailed description, the invention is described withreference to specific exemplary embodiments thereof. Variousmodifications and changes may be made thereto without departing from thebroader spirit and scope of the invention as set forth in the claims.The specification and drawings are, accordingly, to be regarded in anillustrative rather than a restrictive sense.

1. A method comprising the steps of: a) receiving a discrete multitone(DMT) modulated upstream data signal at a first clock rate, c1; and b)interpolating the received signal to a second clock rate c2>c1 toprovide a c2 data signal.
 2. The method of claim 1 further comprisingthe step of: c) processing the c2 data signal through a power spectraldensity shaping filter to provide a power-shaped data signal.
 3. Themethod of claim 2 further comprising the steps of: d) interpolating thepower-shaped data signal to a third clock rate c3>c2 to provide a c3data signal; and e) performing a digital-to-analog conversion on the c3data signal.
 4. The method of claim 1 further comprising the step ofpre-processing the DMT modulated upstream data signal to substantiallyeliminate even images.
 5. The method of claim 4 wherein the step ofpre-processing comprises performing a fast Fourier transform (FFT). 6.The method of claim 5 wherein the FFT is at least a double rate FFT. 7.The method of claim 1 wherein c2=1.104 MHz.
 8. The method of claim 3wherein c3=35.328 MHz.
 9. A method comprising the steps of: a) passing acomposite signal containing discrete multitone modulated upstream anddownstream data signals through a hybrid to extract the downstream datasignal; and b) filtering the extracted downstream data signal through ahigh pass filter having a corner frequency, f1.
 10. The method of claim9 further comprising: c) filtering the high pass filtered signal througha low pass filter having a corner frequency f2>f1; and d) converting thetwice filtered downstream data signal to a digital signal.
 11. Themethod of claim 10 further comprising the steps of: e) decimating thedigital signal from a first rate c1 to a second rate c2, wherein c2<c1;and f) filtering the decimated signal with an anti-aliasing low passfilter.
 12. The method of claim 11 wherein c2=8.836 MHz.
 13. The methodof claim 11 further comprising the steps of: g) decimating theanti-aliased signal to a third rate c3; and h) filtering the twicedecimated signal with second high pass filter.
 14. The method of claim13 wherein c3=2.208 MHz.
 15. The method of claim 9 further comprisingthe step of pre-processing the upstream data signal to substantiallyeliminate even images.
 16. The method of claim 15 wherein the step ofpre-processing further comprises performing a fast Fourier transform(FFT) on the received upstream data signal.
 17. The method of claim 14wherein the FFT is at least a double rate FFT.